Apa fungsi UPS ?

UPS berfungsi untuk menjaga agar perangkat elektronis akan tetap mendapatkan daya listrik apabila tiba-tiba arus listrik dari PLN mati. UPS akan otomatis menswitch dengan kecepatan 0,004 detik dari arus listrik ke power batere, sehingga alat eletronik tetap menyala.

UPS banyak sekali dipakai terutama untuk alat-alat eletronik yang perlu menyala terus selama pekerjaan, misalnya PC (personal computer). Arus listrik yang tiba-tiba mati juga dapat mengakibatkan kerusakan di perangkat komputer. Kita tahu bahwa PLN seringkali melakukan pemadaman arus listrik tanpa pemberitahuan lebih dahulu, atau melakukan pemadaman di waktu jam kerja.

Dari karateristik gelombangnya UPS dapat dibagi menjadi 2 bagian, yaitu :

1. UPS Sin wave (sinus), bentuk gelombang seperti listrik PLN

2. UPS step-square/modified sinwave, bentuk gelombang persegi

UPS sinwave

UPS ini memiliki kelebihan dimana pada waktu arus dari PLN mati, maka otomatis UPS ini akan bekerja dengan waktu switch nol detik. Pada kondisi listrik PLN hidup/mati sumber listrik beban tetap didapat dari inverter/battere. Dimana listrik PLN yang bentuknya arus bolak-balik dirubah menjadi arus searah, kemudian dari arus searah dibuat lagi ke arus bolak balik. Dua kali konversi. Keuntungan dari UPS ini adalah gangguan yg ada pada listrik PLN disaring dengan baik, sehingga terbebas dari gangguan.

UPS step-square/modified sinwave

Pada kondisi listrik PLN hidup maka listrik PLN di teruskan ke beban (PC). Bila listrik PLN mati, maka switch akan segera pindah dan inverter/battere bekerja memberikan listrik ke beban/perangkat (ada jeda waktu pindah 3 s/d 4 mili detik = 0,004 detik).

Tentunya UPS sinwave harganya lebih mahal, karena memiliki beberapa keunggulan dibandingkan tipe square wave. Tentunya pilihan UPS tipe mana yang cocok untuk kebutuhan anda tergantung dari budget dan untuk apa kebutuhan UPS nya.

Kalau anda membutuhkan UPS untuk PC - personal computer, maka UPS tipe square wave dengan harga yang cukup ekonomis sudah memadai. Apabila anda ingin memakai UPS ini untuk suatu Server komputer, maka UPS tipe sinwave akan menjadi pilihan yang tepat.

There are many techniques for measuring capacitance. Some of these techniques require a function generator to provide either a sinusoidal or step-function voltage source. The design idea presented here has the advantage of requiring no special excitation source. Instead it relies on a simple test circuit, along with the single-shot capture and measurement capabilities inherent in digital oscilloscopes (DSOs).
The circuit can accurately measure very small capacitances, and also is able to accurately measure capacitances that change as a function of applied voltage. An example of devices which feature voltage-dependent capacitance is a reversed-biased p-n junction, such as the collector-base junction of a bipolar transistor. Another example is a TVS (transient voltage suppressor diode) device.
The test circuit consists of a single npn transistor (Q1) configured in a common-base connection (Fig. 1). U1 is a constant-current source (LM334) in the emitter leg of the transistor. The transistor exhibits very low collector-base capacitance (CCB = 0.32 pF typical). This specification is critical to the design, as the actual voltage swing across the DUT will occur between the collector of the transistor and ground. The base of the transistor is biased with a constant dc voltage equal to ½ of the supply voltage (-VEE). Maximum VCE for the transistor used in this circuit is 12 V, therefore -VEE should be limited to about -22 V maximum.
The circuit works as follows. The LM324 constant-current source (U1) is programmed for a cathode current of 10 µA by the selection of R1 (67mV/R1 = ICATHODE). The capacitance to be measured is connected between circuit ground and the collector of Q1. A low capacitance switch is used to short the collector of Q1 to ground. Please note-I used the sharp edge of a simple clip-lead to manually contact the ground node, making a very low capacitance connection.
A very low capacitance scope probe (FET type) is attached to the circuit as illustrated in Figure 1. A digital oscilloscope is used in single-shot capture mode to capture the falling edge of the voltage waveform, which appears across the DUT after the short is released. The unknown capacitance is charged by the constant current source through Q1. With proper triggering, the entire voltage waveform can be captured.
The programmed current of the constant-current source can be changed, depending on the range of capacitance value to be measured, and is not critical. The selected value of charging current determines the slope of be displayed waveform. The voltage response (slope) can be made arbitrarily slow so that inductances inherent in the circuit won’t affect the measurement.
The displayed results can be analyzed as follows. Because the DUT has been charged with a constant-current, the capacitance of the device is simply :

C = I/(dV/dT)

Both the time and voltage parameters (slope of the captured voltage waveform) can be measured directly from the displayed waveform. Devices with a fixed capacitance will display a linear slope characteristic (up to the saturation voltage of Q1). Devices with capacitance that is voltage-dependent will show a varying slope characteristic. Capacitance can be directly measured at any bias voltage for devices with a voltage-dependent capacitance.
Certain features of typical modern DSOs make these type of measurements particularly convenient. LeCroy oscilloscopes have a measurement feature called “delta-time-between-levels,” which allows a direct measurement and readout of the delta-time between two cursor-selected voltages on any displayed waveform (s. 2,3, and 4).
The waveform displayed in Figure 2 is captured with the measurement circuit alone, without any DUT. Therefore, this is a baseline measurement of the capacitance of the test circuit. This consists of the capacitance (CCB) of Q1, the scope probe, and the parasitic capacitances of the physical test circuit. The measured value (3.3 pF) will be subtracted from subsequent measurements.
Figure 3 displays a waveform obtained when the DUT is a TVS device. Such a device is termed a “low-capacitance-type” TVS. The manufacturer achieves a low capacitance by inserting a high-speed rectifier (with low capacitance) in series with the TVS diode. It can be seen in the displayed result that the capacitance of the device is indeed very low (3.4 pF) when the device is biased with up to 0.5 Volts. However, above this bias voltage, the internal rectifier diode is conducting, and the capacitance of the TVS device now dominates.
Figure 4 displays two waveforms. Trace 2 is the entire captured waveform showing TVS characteristics from 0 V on up to its breakdown voltage with a bias current of 10 µA (this is a 3-V TVS device). The expanded trace (A) is an expansion of the region from 0 V to approximately-0.5 V (the region of low capacitance). Measurement on this expanded trace yields the capacitance value of 3.4 pF.
This is a convenient technique which uses a simple, small, and portable circuit to measure voltage-dependent capacitance characteristics. This circuit also has been used to measure the parasitic capacitances at input connectors and other areas of pc boards, which could not easily be driven by sinusoidal voltage sources or connected to test instruments for direct measurement.

Among the techniques available to measure airspeed, thermal anemometry has the virtues of simplicity and easy miniaturization. Such anemometers use the relationship between airspeed and power dissipated by a heated sensor known as King’s Law. One good approximation to King’s Law is :

S = A *[(P-D) / (TS - TA)]2

Where :
S = airspeed
A = full-scale calibration
constant
P = power dissipated by
the airspeed sensor
D = “still-air” (S=0)
power dissipation
TS = temperature of the
airspeed sensor
TA = ambient temperature

Two practical problems of thermal airspeed sensors are apparent from this equation. First, the accuracy of the airspeed measurement obviously depends on stability of the (TS - TA) term. This means that either the TS and TA measurements must track very closely, or the (TS - TA) differential must be made large enough to swamp the drift caused by ambient temperature excursions.
Accurate temperature measurement isn’t easy, so the brute-force route usually is chosen, and the sensor is kept good and hot. The penalty of this strategy is power consumption on the order of 1 W, making portable operation problematic.
Also, the second-order exponent makes the raw sensor output nonlinear with airspeed. Therefore, thermal anemometers typically need some provision for measurement linearization.
Figure 1’s circuit utilizes the venerable LM334 temperature sensor to minimize both headaches. LM334s generate a proportional-to-absolute-temperature (PTAT) voltage of ?214µ V/° Kelvin. Therefore, if a constant (VS - VA) voltage differential is maintained, a constant (TS - TA) temperature differential will result. Figure 1’s arrangement of R2, R3, and Q1 provide a stable voltage difference (V1) in the range of 0 to 4 mV to be added to VA.
Op-amp A1 adjusts V2 to maintain VS = VA + V1 and, thereby, TS = TA + V1/214µV. this works because the power dissipation of ambient-sensor T1 is about 100 µW and is, therefore, too little to significantly heat the sensor (LM334s in TO-92 packager have a still-air dissipation constant of 5.6mW/°C). Airflow-sensor T2’s dissipation, however, is much larger: P = 1.06 * (12 – V2) * V2/R4, making T2 heat up when A2’s output slews positive, taking V2 with it. As V2 swings from 0 to 5 V, P goes from 0 to 74 mW . Depending on air speed, this power range is sufficient to maintain a (TS - TA) differential (as set by R2) of 4 to 13° Kelvin. V2 is buffered and zero corrected by A2 and becomes the 0-to-5-V airspeed output signal.
But what about measurement linearization? As illustrated in Figure 2, the inherent quadratic relationship between P and V2 does a reasonable job of canceling King’s Law nonlinearity and results in less than 5% error over more than half of the zero/full-scale range of airspeeds. Anemometers zero/full-scale calibration is straightforward and interaction-free if done in the right sequence. T2 should first be exposed to air flow corresponding to the desired full-scale airspeed and R2 adjusted for VO = 5V. T2 then is placed in calm air and R6 adjusted for VO = 0.
C1 and Q2 provide protection against feedback-loop oscillation and latchup. The 12-V supply should be well-regulated for good circuit stability. Total power consumption depends on airspeed, but never exceeds 144 mW ( 12V @ 12 mA ). This figure is easily six times less than the requirements of comparable performance sensors, especially anemometers that use sturdy plastic sensors instead of fragile metallic filaments. Response is fairly quick (less than 2 seconds) due to the constant-temperature operation of T2.

Generating sine waves with controlled frequencies over a wide range is difficult when using RC or LC sinusoidal oscillators. However, this performance can be simply created using a wideband digital square-wave oscillator, a counter, and a weighted summing network
Using the circuit shown, a sinusoidal output signal with a 100,000,000+:1 frequency range from about 1 MHz to under 0.01 Hz can be obtained without need for any low-pass or high-pass filters. The circuit consists of two parts. The first part is a counter IC with a controlled inverter (IC2) that sequences the switching of input resistor of the second part-a summing amplifier (IC3). The EXOR gates are used to invert signals from four of IC1 counter’s out-puts (q0-q3), depending on logical value at the fifth counter output (Q4). This operation creates the positive and negative halves of the sine waveform. Each of this halves consists of 24 = 16 parts.
The logical values at IC1’s Q0 – Q4 outputs produce weighted symmetrical currents at the summing junction of IC3. The amplifier adds all four weighted currents and generates an output signal with the desired sinusoidal waveform.
Every period of the output signal needs 16 * 2 = 32 periods of input signal, i.e. the frequency of input clock signal must be 32 times higher than the desired frequency of output analog signal :

fOUT-ANALOG = fIN-DIGITAL/32

By changing the values of resistors R1-R4, other waveform can be produced

Of the many starting points for voltage-to-frequency converter (VFC) design, one of the golden oldies is the classic diode-capacitor charge pump. An example of this fundamental circuit is represented by D1, D2, C1, and C2 (see the figure). Analysis of this simple topology reveals that each cycle of A1’s 4-V p-p square wave output will inject a charge onto C5 given by :

Q- = -(C1 + C2 + CS)[(C1 + C2) / (C1 + C2 + CS) – 2Vd]

Where :
V = the peak-to-peak pump-drive voltage generated by A1
CS = stray capacitance at the Dn/Cn common node including diode junction
capacitance
Vd = diode forward voltage drop

An obvious snag with this scheme is that the need to cope with temperature dependence of Vd (approximately – 2 mV/°C) inevitably complicates VFCs that use this basic pump. Elegant compensation circuits exist that work by tweaking V so as to cancel out Vd (pioneered by Bob Pease; and for an interesting twist by Jim Williams, see Linear Technology’s LT1495 data sheet) through relationships like : V = VREF + 2 Vd. But these methods sometimes run into trouble, particularly in micro power applications where the need to make Q dinky (thereby minimizing current consumption) runs afoul of Vs >>0.
The figure illustrates a different Vd fix. In this circuit, D3, D4, C3, and C4 work together to make a compensatory charge pulse for each A1 output cycle :

Q+ = (C3 + C4 + CS)[(V * C3) / (C3 + C4 + CS) – 2Vd]

The pay off is that, if we assume C1 = C4 and C2 = C3, and equality of Vds and Css, then each full cycle of A1 will inject onto C5 a net charge pulse of :

(Q+) + (Q-) = (C1 + C2 + CS){V[(C2 – (C1 + C2)]/(C1 + C2 + CS) + 2Vd - 2Vd }
= V[C2 – (C1 + C2)] = -V*C1

Not only do we get compensation for the bothersome Vds, but the effects of stray capacitance also get rubbed out.
The rest of the figure uses the new self-compensating pump to close a feedback loop around A1so that input currents are balanced by : If = -F0 *V*C1 = 0 to - 1µA as Vin
Goes from 0 to +2.5 V and F0 goes from 0 to 10 kHz. A2 serves to develop a stable drive source V from THE ltc1440 1.2-V internal reference and will do so for supply voltages from 4.5 to 36 V. A3 is a startup circuit that restores oscillation of the A1 charge pump, ac-coupled feedback loop if lockup occurs from, say, input over range.
Overall temperature coefficient of the converter depends on matching of all pump capacitances, including circuit board layout contribution to Cs parasitics. Just ±5% tolerance is good enough to reduce the charge-pump temperature coefficient to approximately 50 ppm/°C. Converter linearity is ±0.03% and current draw is an unexcelled 6.5 to 9 µA @V+ = 5V as F0 goes from 0 to 10kHz.

Adding several components to a simple first-order; low-pass filter; helps to create a different yet handy filter. The circuit shown in Figure 1 combines a low-pass filter (R2, C1, A1) with a bidirectional diode clipping network (R1, D1, D2). The result is a filter that will limit the maximum slope (not frequency) it passes.
Typical uses for this circuit are shown in Figure 2a-2d. in general, it’s used to create ramps from step voltages, generate triangle/trapezoid waveforms from square waves, remove unwanted fast components (noise/transient) from any signal, or limit the maximum rate of change of any signal.
Here’s how it works. Whenever the input voltage VIN differs from the output voltage VOUT by one forward diode drop or more, one of the diodes will turn on ( D1 when VIN > VOUT and D2 when VIN < VOUT ). When this happens, the voltage across R2 is held fairly constant (because the voltage at the “+” input and the output of A1 are equal) at one forward diode drop.
With a fixed voltage across R2 and, therefore, a constant current through it, the capacitor C1 charges linearly in stead of exponentially. The maximum slope (V/T) that the circuit will pass is equal to the VF of the diodes used divided by R2C1 (maximum slope (V/T) = VF /R2C1). This assumes R2 >>R1. No matter how quickly the input voltage changes, the output will never change any faster than the limit set by R2C1. any signal or part of a signal with a slope less than this limit simply passes through the circuit unaffected.
VIN should be driven by a low impedance source. Resistor R1 limits the current through D1 or D2 when they conduct. Typically R1 is 1k-10k. its value should be kept as small as practical and depends on the drive capabilities of A1 and the op-amp or other device driving VIN. R2’s resistance should be much greater than that of R1 to swamp out its contribution to the circuit’s R2C1 time constant. R2 and C1 form a low-pass filter and A1 buffers it and provides a low impedance path for D1 or D2 when in conduction. For the best performance, D1 and D2 should be low VF (Schottky) types, although other diode types (1N914, 1N4148, etc) will work satisfactorily.
When a square wave or step voltage is slope-limited by this circuit, a slight rounding will be seen at the top and bottom of the output waveform. This is due to the loss of overhead voltage (needed for diode conduction) that occurs when the capacitor has charged within one diode drop of the input’s peak voltage. This rounding is minimized by using low VF diodes, keeping R1 as small as possible, and by using the largest amplitude input waveform your supply voltage will allow.
I originally came up with this circuit while designing a servo control loop. I needed a simple way to limit the rate of change of the servo’s output signal. It also can be used to soft start lights, create smooth motor speed transitions, filter a signal by its slope instead of its frequency, tame ill-behaved servo circuits, slow square wave transitions (without excessive rounding), and so on. Unlike integrator-based circuits this circuit works with single-ended or bidirectional supplies.

Spesification :
1. WM 01

Spesification :
1. BF 02
2. BF 03

Product Category: 19″ RACK ENCLOSURES
Product Series : ACCESORIS
Product Name : Ventilation Panel
Brochure : Cabinet Rack.pdf

Spesification :
1. VP 01
2. VP 02
3. VP 03
4. VP 04
5. VP 05

Spesification :
1. CT 42
2. CT 45